Abstrict A vortex flow meter produces Karman's vortices in a fluid to be
measured and a piezoelectric sensor develops a vortex signal therefrom
which is proportional to the flow velocity of the fluid. The output
signal of the sensor is applied to a charge amplifier and a low
pass filter coupled thereto which functions to a predetermined signal
amplitude for limiting low and high frequency noise and stabilizing
the gain of the system within the vortex frequency measuring range.
The output signal is converted by a logic circuit to a pulse signal
of vortex frequency and then converted to a DC signal for transmission
to an indicator or a receiver.
Claims We claim:
1. In a vortex flow meter having a vortex generator for producing
Karman's vortices proportional to the flow velocity in a fluid to
be measured and a piezoelectric sensor for detecting a vortex signal
based on the Karman's vortices as a change in the quantity of electric
charges comprising in combination:
(a) a charge amplifier coupler to said piezoelectric sensor for
converting the alternating charge output of said sensor into an
AC signal voltage;
(b) a filter circuit having a low-pass characteristic coupled to
said charge amplifier;
(c) means coupled to said filter circuit for inactivating the low-pass
filter characteristics of said filter circuit when said signal voltage
exceeds a predetermined level;
(d) means coupled to said filter circuit for converting said signal
voltage into a pulse signal;
(e) a time constant circuit having said pulse signal applied thereto;
(f) a logic circuit having said pulse signal and the other of said
time constant circuit applied thereto functioning as an exclusive
OR gate for producing an output pulse signal whose duty ratio corresponds
to the vortex frequency; and
(g) an output circuit coupled to said logic circuit for converting
the pulse output of said logic circuit into a DC signal representing
the flow velocity or quantity of the fluid to be measured.
2. In the vortex flow meter as defined in claim 1 wherein said
filter circuit of low-pass characteristic combined with the charge
amplifier is an active filter, whose cutoff frequency is selected
to be lower than that of the charge amplifier so as to compensate
the low-range characteristic of said amplifier.
3. In the vortex flow meter as defined in claim 2 wherein said
active filter is a single feedback type low-pass filter comprising
an operational amplifier and an impedance element.
4. In the vortex flow meter as defined in claim 1 wherein said
filter circuit includes an operational amplifier, a series circuit
comprising a Zener diode and a resistor coupled between the input
and output terminals of said operational amplifier for inactivating
the low-pass filter characteristic of said filter circuit when the
output of said operational amplifier exceeds a predetermined level.
5. In the vortex flow meter as defined in claim 1 wherein two
piezoelectric sensors are provided integrally with the vortex generator
so as to detect a vortex signal based on the Karman's vortices as
a change in the quantities of mutually opposite-phase electric charges,
said two sensors being positioned on the two sides of the neutral
axis of the vortex generator.
6. In the vortex flow meter as defined in claim 1 wherein two
piezoelectric sensors are provided integrally with a pressure receiver
located in the downstream of the vortex generator, so as to detect
a vortex signal based on the Karman's vortices as a change in the
quantities of mutually opposite-phase electric charges, said two
sensors being positioned on the two sides of the neutral axis of
the pressure receiver.
7. In the vortex flow meter as defined in claim 1 wherein said
logic circuit is composed of a CMOS.
8. In the vortex flow meter as defined in claim 1 wherein said
time constant circuit has a variable resistor for changing to time
constant thereof, thereby providing a span adjustment for said vortex
flow meter.
9. A vortex flow meter for measuring the flow velocity of a fluid
comprising in combination:
(a) a vortex generator for producing Karman's vortices proportional
to the flow velocity of a fluid to be measured;
(b) a piezoelectric sensor for producing a vortex signal based
on the Karman's vortices as a change in the quantity of electric
charges;
(c) a charge amplifier coupled to said piezoelectric sensor for
converting the alternating charge output of said sensor into an
AC voltage;
(d) a low-pass filter coupled to said charge amplifier;
(e) means coupled to said low-pass filter for inactivating the
filter characteristic of said low-pass filter when said signal voltage
exceeds a predetermined amplitude level;
(f) means coupled to said low-pass filter for converting said signal
voltage into a pulse signal having a vortex frequency;
(g) a time constant circuit;
(h) means for applying said pulse signal to said time constant
circuit;
(i) a logic circuit having said pulse signal and the output of
said time constant circuit applied thereto functioning as an exclusive
OR gate for producing a pulse cycle whose duty ratio corresponds
to the vortex frequency;
(j) an output amplifier having a pair of differential input terminals;
(k) a resistor for coupling the output of said logic circuit to
one of said differential input terminals of said output amplifier;
(l) a pair of voltage dividers and a feedback voltage;
(m) means for coupling one of said voltage dividers to said one
of said differential input terminals;
(n) means for coupling said feedback voltage from the other of
said voltage dividers to the other differential input terminal of
said output amplifier;
(o) an output transistor driven by said output amplifier;
(p) a pair of transmission lines having a load connected in series
to a DC power source on the receiving side thereof, being coupled
to said output transistor for controlling the current through said
load; and
(q) a feedback transistor for generating said feedback voltage
corresponding to the output current flowing in said load, whereby
said current flowing in said load on the receiving side of said
transmission lines represents the flow velocity or quantity of the
fluid to be measured.
10. A three wire vortex flow meter for measuring the flow velocity
of a fluid comprising in combination:
(a) a vortex generator for producing Karman's vortices proportional
to the flow velocity of a fluid to be measured;
(b) a piezoelectric sensor for detecting a vortex signal based
on the Karman's vortices as a change in the quantity of electric
charges;
(c) a charge amplifier coupled to said sensor for converting the
alternating charge output of said sensor to an AC signal voltage;
(d) a filter circuit having a low-pass characteristic coupled to
said charge amplifier;
(e) means coupled to said filter circuit for inactivating said
low-pass filter characteristic therefrom when said signal voltage
exceeds a predetermined level;
(f) means coupled to said filter circuit for converting said signal
voltage into a pulse signal corresponding to the vortex frequency;
and
(g) means including three transmission lines for transmitting said
pulse signal to a receiver.
Description BACKGROUND OF THE INVENTION
The present invention relates to a vortex flow meter for measuring
the flow velocity or quantity of a fluid by utilizing Karman's vortices.
More particularly, this invention relates to such a meter equipped
with a piezooelectric sensor and a charge amplifier.
It is known that when an object is placed in a fluid, vortices
are generated on the sides of the object alternately and regularly
to form a flow of vortex street in the downstream. This is referred
to as a Karman's vortex street, and the number of vortices (vortex
frequency) generated during a unit time is proportional to the flow
velocity of the fluid.
The vortex flow meter has a vortex generator disposed in a duct
carrying the fluid to be measured, and Karman's vortices proportional
to the flow velocity are generated therefrom. These vortices are
detected by means of a sensor such as a thermosensitive element
or a piezoelectric element to produce an electrical signal corresponding
to the flow velocity or quantity of the fluid. One vortex flow meter
employing a piezoelectric element to detect the fluid vibration
as a change in an AC voltage is disclosed in the U.S. Pat. No. 3948098.
The piezoelectric sensor is further capable of detecting the fluid
vibration as a change in the electric charge quantity. In this case,
the charge quantity obtained from the piezoelectric sensor is converted
into a voltage signal by means of a charge amplifier, of which the
cutoff frequency normally is selected to be below the minimum value
(1 Hz) of the vortex frequency to be measured so that a satisfactory
response characteristic is attained within the vortex frequency
range (approximately from 1 Hz to 120 Hz when the fluid to be measured
is a liquid). For ensuring excellent low-range characteristics in
the charge amplifier, it is necessary to select a large value for
the time constant of a resistor-capacitor feedback circuit for the
amplifier. However, the sensitivity of the amplifier is dependent
on the value of the capacitor, which should be reduced to achieve
a high sensitivity. Therefore, increasing the resistance is required
to attain a large time constant. For example, when setting the cutoff
frequency of the charge amplifier to 1 Hz, the resistance required
becomes extremely high, e.g. above 1000 megohms. A problem arises
with respect to the reliability in the resistance of any value exceeding
1000 megohms, in addition to a disadvantage of high production
cost. Thus, practical use has not yet been achieved for a vortex
flow meter of the type that performs signal processing after detecting
the fluid vibration as a change in the electric charge quantity
by a piezoelectric sensor.
A noise component referred to as fluctuation of frequencies lower
than the vortex frequencies (ranging from 1 to 120 Hz) is superposed
on the vortex signal. The noise frequency becomes higher with increasing
vortex frequency, and its magnitude also increases with the frequency.
Moreover, when detecting the vortex signal by a piezoelectric sensor,
detection is affected by noise such as duct vibration caused by
a pump or the like. The noise frequencies resulting from duct vibration
are within a range from several ten hertz to several hundred hertz,
and the magnitude thereof increases generally in proportion to the
frequency. In a vortex flow meter, it is desired that any harmful
effect of such noise components be effectively eliminated to achieve
adequate detection of the vortex signal at a satisfactory signal-to-noise
ratio throughout a wide range of flow velocities.
SUMMARY OF THE INVENTION
It is the principal object of the present invention to provide
a new vortex flow meter having a piezoelectric sensor and a charge
amplifier without the above-described disadvantages.
Another object of the invention is to provide a vortex flow meter
wherein an active filter having low-pass characteristics is combined
with a charge amplifier for compensating the low-range characteristics
of the charge amplifier by the filter as well as effectively eliminating
high-frequency noise above the vortex frequency.
Another object of the invention is to provide a vortex flow meter
wherein the filter characteristics of an active filter is inactivated
when the voltage from a charge amplifier exceeds a predetermined
level, so as to effectively eliminate the harmful effect of the
fluctuation noise component which is superposed on the vortex signal
and has frequencies lower than the vortex frequencies.
A further object of the invention is to provide a two-wire vortex
flow meter which is capable of accurately converting a vortex-frequency
signal of alternating electric charge into a DC signal of, for example,
4 to 20 milliamperes through a pair of transmission lines to a DC
power source and a load located on the receiving side.
In attaining these and other objects and in carrying out this invention
in one illustrative embodiment thereof a vortex flow meter is provided
having a sensor for detecting a vortex signal which is proportional
to the flow velocity of a fluid. The vortex signal is applied to
a charge amplifier and a low-pass filter circuit in which the filter
operates on the vortex signal until the signal exceeds a predetermined
amplitude for eliminating noise components in the vortex signal
having frequencies lower that the vortex frequencies to be measured
as well as high frequency noise components above the vortex frequency.
The vortex frequency signal is then converted to a pulse signal
of vortex frequecy and to a DC signal for transmission to a utilization
means.
BRIEF DESCRIPTION OF THE DRAWINGS
FIG. 1 illustrates the structure of a vortex flow meter embodying
the present invention;
FIG. 2 is an electrical schematic diagram of an embodiment of the
invention;
FIG. 3 illustrates the structure of one form of piezoelectric sensor
used in the invention;
FIG. 4 graphically represents characteristic curves for explaining
the operation of the vortex flow meter of the invention;
FIGS. 5 through 8 show signal waveforms for explaining the operation
of the invention; and
FIGS. 9 through 11 are electrical schematic diagrams of other embodiments
of the invention.
DESCRIPTION OF THE PREFERRED EMBODIMENTS
Referring now to FIG. 1 the instrument includes a detector, generally
indicated at 10 for detecting the vortices produced in a duct 11
by a pillar-shaped vortex generator 12. This generator is positioned
perpendicularly to the duct 11 and is anchored thereto at two ends.
The body 12a of the vortex generator 12 is made of a rigid substance
such as stainless steel and has a trapezoidal or similar cross sectional
shape suitable for generating a Karman's vortex street in the fluid
to be measured and also for stably intensifying the fluid vibration.
The top 12b of the vortex generator 12 is made of a rigid substance
such as stainless steel and has a recess 12c. The top 12b is formed
integrally with the body 12a by welding or the like.
A piezoelectric element 13 formed of lithium niobate (LiNbO.sub.3)
or the like is sealed in the recess 12c of the vortex generator
12 with an insulating material 14 such as glass. The piezoelectric
element 13 is shaped like a disk and is so disposed that its center
coincides with the neutral axis of the vortex generator 12. As illustrated
in FIG. 3 the piezoelectric element 13 is equipped with electrodes
15a, 15b, 15c and 15d located on its obverse and reverse sides symmetrically
at the left and right, wherein a portion between the electrodes
15a and 15b forms a first piezoelectric sensor 16a, while a portion
between the electrodes 15c and 15d forms a second piezoelectric
sensor 16b. The electrodes 15a and 15d are connected to each other,
and similarly the electrodes 15b and 15c are connected to each other
such that the electric charges generated in the first and second
piezoelectric sensors 16a and 16b are coupled differentially by
leadwires 17a and 17b which extend from the electrodes 15a and 15c
via the insulating material 14 and are connected to the converter
20 both electrically and mechanically.
When the fluid to be measured flows in the duct 11 the vortex
generator 12 produces Karman's vortices and receives an alternating
force resulting from the fluid vibration resulting from generation
of the vortices. Upon application of the alternating force to the
vortex generator 12 the stress is changed in mutually opposite
directions therein on the two sides of the neutral axis as illustrated.
The stress change thus caused in the vortex generator 12 is transmitted
to the piezoelectric element 13 through the insulating material
14. Accordingly, in the first and second piezoelectric sensors 16a
and 16b, opposite-phase changes of the electric charge quantities
are produced corresponding to the stress change.
The noise component including duct vibration causes corresponding
vibration of the vortex generator 12 together with the duct 11 since
the generator is made of a rigid substance, so that the vibration
appears in the form of acceleration in the piezoelectric sensors
16a and 16b, and thus the majority thereof causes in-phase changes
of the electric charge quantities. The charge quantities produced
in the piezoelectric sensors 16a and 16b are detected differentially,
thereby doubling the opposite-phase charge quantities based on the
signal. However, the in-phase charge quantities cancel each other
to become sufficiently small, so that the signal obtained between
the leadwires 17a and 17b has a satisfactory signal-to-noise ratio.
The alternating charge q of a vortex frequency f produced between
the leadwires 17a and 17b is applied to a converter 20. The quantity
of the alternating charge q is proportional to the square of the
vortex frequency f.
The alternating force resulting from the fluid vibration of the
Karman's vortices is received by the vortex generator 12 and the
stress caused therein is detected by the piezoelectric sensors located
in the generator 12 so that a remarkably simplified construction
is provided without employing any movable component parts, by using
a solid and durable detector with high sensitivity. Moreover, since
the alternating force resulting from the fluid vibration is received
by the entire vortex generator, the detector is affected least by
the flow velocity distribution of the fluid being measured. Furthermore,
in view of the fact that the sensor is not in direct contact with
the fluid, a suitable anticorrosive material may be selected freely
for the component parts which are in contact with the fluid, and
there is no restriction with respect to the type of coating, thereby
permitting use of a detector with a highly corrosive fluid as well.
In addition to the above, excellent heat resistance is attainable
by employing glass or the like having a high heat-resisting property
for the insulating material 14 to seal the piezoelectric element
13 in the vortex generator 12. Thus, a variety of advantageous features
are achievable including applicability for use in a high-temperature
fluid.
Although the detector 10 of the foregoing embodiment illustrates
the two ends of the vortex generator 12 being fixed to the duct
11 the detector 10 may be fixed on one end and free on the other
end or fixed on one end and supported on the other end thereof.
Fixing means, such as welding, screwing or bolting may be selected
as desired. The piezoelectric element 13 of lithium niobate may
be replaced with a piezoelectric crystal of lithium niobate or quartz,
or a ceramic piezoelectric or pressure-sensing element of zircon
titanate (PZT) or tinanate. In short, any element capable of converting
a force into a charge quantity is usable. The type of insulating
material 14 employed is not limited to glass alone, but may include
other material such as epoxy, ceramic, cement or mica. Any suitable
material which is an electric insulator with chemical stability
capable of transmitting the force produced in the vortex generator
12 to the piezoelectric element with certainty and high sensitivity
may be used. A detector having the above structure is disclosed
and described in U.S. Pat. application Ser. No. 910638 filed on
May 20 1978 and is assigned to the assignee of the present invention.
In an embodiment of this invention the piezoelectric element 13
is shown located in the vortex generator 12. However, the structure
may be modified to provide a pressure receiver to receive an alternating
force resulting from fluid vibration downstream of the vortex generator
12 and separated therefrom. In such modification the piezoelectric
element 13 is sealed in the pressure receiver by the insulating
material 14 such as glass, so that a stress change occurring in
the pressure receiver is detected. Furthermore, the detector 10
is not limited to the type for detecting the stress produced in
the vortex generator or in the pressure receiver as a result of
fluid vibration, and may have various constructions that detect
the fluid vibration by means of a piezoelectric sensor.
The converter 20 of the vortex flow meter comprises a charge amplifier
21 with a filter circuit 22. The amplified signal is applied to
a comparator 23 and then to a logic circuit 24 which includes a
time constant circuit 25. The resulting signal is coupled to an
output amplifier 26 having an output transistor 27 and supplied
by a constant voltage circuit 28. The output transistor 27 is connected
in series with a feedback resistor R.sub.f to form a series circuit,
which is further connected through a pair of transmission lines
30 to a series circuit of a DC power source 40 and a load 50 located
on the receiving side, so that the output current I.sub.o controlled
by the output transistor 27 flows in the feedback resistor R.sub.f
and the load 50. The constant voltage circuit 28 is connected in
parallel to the output transistor 27 and consists of a constant
current circuit J having a field-effect transistor Q.sub.1 and
a zener diode DZ.sub.1 connected in series thereto through a resistor
R.sub.20. A stabilized voltage Es is generated across the diode
DZ.sub.1.
Voltage Es is applied to the power terminal of logic circuit 24
and also to one input terminal (-) of an operational amplifier OP.sub.4
the output amplifier 26 through voltage-dividing resistors R.sub.16
and R.sub.17. Meanwhile, the voltage Es is superposed on a feedback
voltage IoRf produced across the feedback resistor R.sub.f and then
is applied to the other input terminal (+) of the operational amplifier
OP.sub.4 through voltage-dividing resistors R.sub.18 and R.sub.19.
Also, the voltage obtained across the series circuit of the resistor
R.sub.20 and the zener diode DZ.sub.1 is fed as a line voltage V
through a transistor Q.sub.2 to operational amplifiers OP.sub.1
OP.sub.2 OP.sub.3 and OP.sub.4 in the charge amplifier 21 filter
circuit 22 comparator 23 and output amplifier 26.
Charge amplifier 21 includes a capacitor C.sub.1 and a resistor
R.sub.1 connected to the feedback circuit of operational amplifiers
OP.sub.1 and OP.sub.2 and leadwires 17a and 17b of the detector
10 are connected between the input terminals of the charge amplifier
21 through a coupling capacitor C.sub.o. The line voltage V is applied
between the power terminals of OP.sub.1 and also the voltage divided
by resistors R.sub.2 and R.sub.3 is applied to the input terminal
(+) of OP.sub.1 through a resistor R.sub.4. The charge amplifier
21 receives the alternating charge quantity q at the input terminals
thereof from the detector 10 and serves to convert the input into
an AC voltage e.sub.1. Since a variation in the capacities of the
leadwires 17a and 17b causes no influence on the output voltage,
the charge amplifier 21 prevents deterioriation of the signal-to-noise
ratio that could result from a touching of the leadwires 17a and
17b. Consequently, it becomes possible to make the leadwires sufficiently
long, thereby facilitating the installation of the converter 20
at a location spaced away from the detector 10. As the piezoelectric
element 13 is sealed in the detector 10 with the insulating material
14 such as glass, the insulation resistance of the piezoelectric
sensor is normally of a considerably large value. When the detector
10 is used in high temperature application, there occurs a sharp
decrease in the insulation resistance of the piezoelectric element
13 of niobate lithium and also in the insulation resistance of the
insulating material 14 such as glass, thereby reducing the potential
at the input terminal (-) of the operational amplifier OP.sub.1
causing saturation of the output e.sub.1 of OP.sub.1. However, the
charge from the detector 10 is received by the charge amplifier
21 through a coupling capacitor C.sub.o prohibiting direct current
from flowing toward the detector 10 thereby preventing a vibration
of the operating point of the amplifier 21 based on a decrease in
the insulation resistance of the detector 10. The capacitance of
the coupling capacitor C.sub.o is selectively set to a value sufficiently
larger than the equivalent capacitance C.sub.s of the piezoelectric
element 13 so as not to affect the gain of the charge amplifier
21 substantially. Therefore, even when using the detector 10 in
high temperature applications, the operation of the charge amplifier
remains stable without any reduction of its sensitivity. In case
static charge is accumulated in the piezoelectric sensor, a discharging
resistor R.sub.o may be connected in parallel thereto as required.
The capacitor C.sub.2 connected between the power terminal and the
input terminal (+) of OP.sub.1 served to prevent the application
of signals from the comparator 23 and so forth to the power supply
line.
The filter circuit 22 is illustrated as a single feedback type
low-pass filter having an operational amplifier OP.sub.2 an input
impedance circuit 22a consisting of resistors R.sub.5 R.sub.6 and
a capacitor C.sub.4 connected to the input circuit of OP.sub.2
and a feedback impedance circuit 22b consisting of resistors R.sub.7
R.sub.8 and capacitors C.sub.5 C.sub.6 connected to the feedback
circuit of OP.sub.2. The output terminal of the charge amplifier
21 is connected to the input terminal of the filter circuit 22 through
a coupling capacitor C.sub.3. The alternating charge quantity q
obtained from the detector 10 increases in proportion to the square
of the vortex frequency f. Noise, such as duct vibration, mostly
consists of frequencies higher than the vortex frequencies ranging
from 1 to 120 Hz. By providing the filter circuit 22 gain of the
entire circuit consisting of the piezoelectric element 13 the charge
amplifier 21 and the filter circuit 22 is maintained substantially
constant within a vortex-frequency measuring range, and the noise
left unremoved by the detector 10 (such as extremely great duct
vibration or noise resulting from unbalance caused between the first
and second piezoelectric sensors 16a and 16b in the manufacturing
process) is attenuated completely. Furthermore, AC voltage e.sub.2
is produced having a satisfactory signal-to-noise ratio. Also, the
low-range characteristics of the charge amplifier 21 can be compensated
by using an active filter including the operational amplifier OP.sub.2
as illustrated. That is, for attaining excellent low-range characteristics
in the charge amplifier 21 it is necessary to establish a large
time constant determined by the resistor R.sub.1 and the capacitor
C.sub.1. Since, the sensitivity of the charge amplifier is dependent
on the value of capacitor C.sub.1 which should be small to achieve
a high sensitivity the value of resistor R.sub.1 must be large to
increase the time constant. For example, when lowering the cutoff
frequency of the charge amplifier below the minimum vortex frequency
(1 Hz), the required magnitude of resistor R.sub.1 is extremely
high, e.g. above 1000 megohms. A problem arises with respect to
reliability of a resistor having a value exceeding 1000 megohms,
in addition to the disadvantage of production cost. In the embodiment
of this invention, therefore, the cutoff frequency fc=(1/2.pi.C.sub.1
R.sub.1) of the charge amplifier 21 is selected to be higher (e.g.
30 Hz) than the minimum vortex frequency, and the value of the capacitor
C.sub.1 is selected to be greater than the equivalent capacitance
C.sub.s of the piezoelectric element 13 so as to set the gain of
the charge amplifier to C.sub.s /C.sub.1 <1. The value of the
resistor R.sub.1 is selected to be sufficiently small (e.g. 50 megohms)
at the sacrifice of the low-range characteristics as shown by a
dotted line (a) in FIG. 4. The cutoff frequency fa of the active
filter 22 is selected to be in the vicinity of the minimum vortex
frequency as shown by a dotted line (b) in FIG. 4 with the gain
being set to a sufficiently large value. Accordingly, the characteristics
of the entire circuit consisting of the piezoelectric element 13
the charge amplifier 21 and the active filter 22 becomes that as
illustrated by a solid line in FIG. 4 wherein the low-range characteristics
of the charge amplifier 21 are compensated by the active filter
22 to attain a desired gain. Thus, the gain of the entire circuit
is rendered constant within a frequency range between fa and fc,
and a reduction of, for example, -20 dB/dec is achieved at any frequency
above fc. For this reason, the noise of any frequency higher than
the measuring frequency range can be fully attenuated with respect
to its amplitude, hence attaining an improved signal-to-noise ratio.
By combining the active filter 22 with the charge amplifier 21 in
this manner, both the low-range characteristics and the sensitivity
of the charge amplifier 21 are compensated by the active filter
22 so that the magnitude of the resistor R.sub.1 in the charge
amplifier 21 can be greatly reduced. When it is desired to further
reduce the value of the resistor R.sub.1 in the charge amplifier,
the output voltage e.sub.1 of OP.sub.1 may be divided and applied
to the resistor R.sub.1 to accomplish a reduction proportional to
the voltage division ratio.
When the fluid to be measured moves at a low velocity, the waveform
of the AC voltage e.sub.1 produced at the output of the charge amplifier
21 includes a high-frequency noise such as duct vibration superposed
thereon as shown in FIG. 5. And when the fluid moves at a high velocity,
the waveform includes beat signals with low-frequency fluctuations
producing a voltage e.sub.1 through the low-pass filter 22 as shown
in FIG. 6 indicating that high-frequency noise is removed at the
low-velocity flow of the fluid to attain a good signal-to-noise
ratio, but that the output in high velocity flow is saturated due
to the low-frequency noise producing a waveform in which high-frequency
signal components are not distinguishable. This is based on the
fact that the signal components have high frequencies in the high
velocity flow and are therefore attenuated in the filter circuit
22 while the low-frequency fluctuation components of the noise
become large in the high velocity flow.
In the present invention, a series circuit comprising a zener diode
DZ.sub.2 and a resistor R.sub.21 is connected between the input
and output terminals of the operational amplifier OP.sub.2 of the
filter circuit 22. When the output e.sub.2 of OP.sub.2 exceeds the
zener voltage VZ, feedback is applied to OP.sub.2 through the series
circuit of DZ.sub.2 and R.sub.21 for inactivating the filter characteristics
while providing amplitude-limiting characteristics. Accordingly,
by selectively setting the zener voltage VZ to a value equivalent
to the output voltage e.sub.2 of OP.sub.2 in which the vortex frequency
is in the vicinity of 6 to 10 Hz, the waveform of the output e.sub.2
is produced which is illustrated in FIG. 7. As will be apparent
from FIG. 7 the high-frequency noise components of the waveform
are eliminated when e.sub.2 is below VZ during low velocity flow,
while the waveform is not affected by the low-frequency noise components
when e.sub.2 is above VZ during high velocity flow, so that the
signal obtained is at a satisfactory signal-to-noise ratio during
both the low and high velocity flow.
Inactivation of the low-pass filter 22 may be achieved by the zener
diode DZ.sub.2 alone even with omission of the resistor R.sub.21
in the series circuit. However, this omission causes a slight distortion
in the output waveform. Alternate means for inactivating the filter
characteristics of the filter circuit 22 may be provided in the
form of a switch connected in parallel with circuit 22 which is
turned on when the output of the filter circuit 22 or the charge
amplifier 21 related to the signal voltage exceeds a preset value.
However, the embodiment employing a zener diode is more advantageous
because of simplicity.
The comparator 23 is illustrated as a Schmitt trigger having an
operational amplifier OP.sub.3 and a resistor R.sub.9 for providing
positive feedback for amplifier OP.sub.3. The output terminal of
the filter circuit 22 is connected to the input terminal of the
comparator 23 through a coupling capacitor C.sub.7. The line voltage
V is applied between the power terminals of OP.sub.3. A voltage
obtained through the division of line voltage V by a voltage divider
of resistors R.sub.10 R.sub.11 and R.sub.12 is fed to the input
terminals (-) and (+) of amplifier OP.sub.3 via resistors R.sub.13
and R.sub.14 respectively, thereby providing a preset trigger-level
for the Schmitt trigger. The values of the resistors R.sub.13 and
R.sub.14 are selected to be sufficiently greater than those of the
resistors R.sub.10 R.sub.11 and R.sub.12 and a capacitor C.sub.8
is connected in parallel to the resistor R.sub.12. The comparator
23 receives the AC voltage e.sub.2 through the filter circuit 22
having a vortex frequency f as shown in FIG. 8(a) and serves to
convert the voltage e.sub.2 into a pulse signal P.sub.1 of a fixed
level as shown in FIG. 8(b).
The logic circuit 24 is illustrated as a C-MOS gate circuit having
two exclusive OR gates G.sub.1 and G.sub.2 wherein an output terminal
of the comparator 23 is connected to one input terminal of G.sub.1
while the other input terminal G.sub.1 is connected to the reference
side of a fixed voltage Es applied to G.sub.1 and G.sub.2. The output
terminal G.sub.1 is connected to one input terminal of G.sub.2 and
also to a time constant circuit 25 whose output terminal is connected
to the other input terminal of G.sub.2. The exclusive OR gate produces
an output "1" when the states of signals fed to the two
input terminals thereof are "1" and "0" or "0"
and "1", or produces an output "0" when the
input signal states are "1" and "1" or "0"
and "0". With a threshold voltage being set to E.sub.T
as shown in FIG. 8 for discriminating between "1" and
"0", a pulse signal P.sub.2 of FIG. 8(c) having a fixed
amplitude Es is produced at the output terminal of G.sub.1. The
signal P.sub.2 is fed to the time constant circuit 25 consisting
of a resistor R.sub.10 and a capacitor C.sub.9 and is converted
into a signal P.sub.3 as shown in FIG. 8(d) which rises and falls
with a delay according to the time constant C.sub.9 .multidot.R.sub.10
of the circuit 25. This time constant is selected to be sufficiently
smaller than the pulse width of the signal P.sub.1. As the signal
P.sub.3 is fed to G.sub.2 together with the signal P.sub.2 a pulse
signal P.sub.o shown in FIG. 8(e) is produced at the output terminal
of G.sub.2 during a time t.sub.1 required for the output of the
time constant circuit 25 to reach E.sub.T from zero and also during
a time t.sub.2 required for the said output to reach E.sub.T from
Es. The amplitude Es of the signal P.sub.o is constant, and its
pulse width t=(t.sub.1 +t.sub.2) in each period T is also constant.
The pulse width t is maintained substantially at a fixed value in
spite of any variation of the threshold voltage E.sub.T resulting
from temperature fluctuation. That is, the characteristic attained
is such that the threshold voltage E.sub.T of the C-MOS becomes
a half of the line voltage Es near normal temperature. Since the
amplitude of the signal fed to the input of G.sub.2 is also constant,
when E.sub.T drops to decrease t.sub.1 in the pulse width t=(t.sub.1
+t.sub.2) of the signal P.sub.o, t.sub.2 increases by an equivalent
amount to provide compensation. On the other hand, when E.sub.T
rises to increase t.sub.1 then t.sub.2 decreases by an equivalent
amount to provide compensation. The pulse width t is easily adjustable
by changing the time constant C.sub.9 .multidot.R.sub.10 by means
of the variable resistor R.sub.10. Consequently, the output pulse
signal P.sub.o of the logic circuit 24 has a duty ratio t/T exactly
proportional to the vortex frequency f. Signal P.sub.o is fed to
the input terminal (-) of the output amplifier 26 through a resistor
R.sub.15. The use of a C-MOS as the logic circuit 24 provides a
circuit substantially free from the harmful influences of temperature
fluctuations and also minimizes power consumption.
As shown in FIG. 9 the output P.sub.3 of the time constant circuit
25 may be fed to an exclusive OR gate G.sub.3 through G.sub.2. In
this case, a gate G.sub.4 is provided for inverting the output P.sub.o
and G.sub.2 since a constant voltage circuit 28 is provided on
the high-voltage side and a feedback resistor R.sub.f is provided
on the low-voltage side in FIG. 9. Although the gates G.sub.1 and
G.sub.3 are illustrated to function as exclusive OR gates, each
may function as an OR gate or a NAND gate.
Referring again to FIG. 2 the output amplifier 26 consists of
an operational amplifier OP.sub.4 whose input terminal (-) is connected
to the output terminal of the logic circuit 24 through a resistor
R.sub.15. The line voltage V is applied between the power terminals
of OP.sub.4 and a voltage obtained through division of the fixed
voltage Es by resistors R.sub.16 and R.sub.17 is fed to one input
terminal (-) of OP.sub.4 while the other input terminal thereof
receives a voltage obtained through division of the superposed fixed
voltage Es and feedback voltage IoRf by resistors R.sub.18 and R.sub.19.
A capacitor C.sub.10 for smoothing the pulse signal P.sub.o is connected
to the resistor R.sub.15. Accordingly, the potential Ea at the input
terminal (-) of OP.sub.4 and the voltage Eb at the input terminal
(+) thereof are represented by the following equations. ##EQU1##
where Rf<<R.sub.18 R.sub.19 The gain of OP.sub.4 is sufficiently
large, and an output transistor 27 is so driven that Ea and Eb become
equal to each other to control the output current Io. Accordingly,
the current Io is expressed as ##EQU2## in Equation (3), since each
resistance value, the pulse width t and the voltage Es are constant,
the output current Io corresponds accurately to the vortex frequency
f=(1/T) or the flow velocity of the fluid. The output current Io
thus obtained is transmitted through a pair of transmission lines
30 to a load 50 located at a receiver on the receiving side.
The zero point of the output control Io may be made adjustable
making resistor R.sub.19 variable (as shown in FIG. 9), and the
span of the meter is adustable by the variable resistor R.sub.10.
It is easy, therefore, to obtain a desired output current Io ranging,
for example, from DC 4 to 20 milliamperes against a vortex-frequency
change ranging from 0 to 100 percent. And there is a particular
advantage that the zero point is not drifted by span adjustment,
since such adjustment is executed by changing the pulse width t
of the signal P.sub.o by means of the variable resistor R.sub.10
in the time constant circuit 25.
Although the aforesaid embodiment of this invention illustrates
the converter 20 being mounted on the detector 10 it is also possible
to mount only the front stage 20a of the converter 20 consisting
of the charge amplifier 21 the filter circuit 22 and the comparator
23 on the detector 10. When disposing the converter 20 by separating
the same into its front stage 20a and rear stage 20b, a malfunction
of the logic circuit 24 may occur due to the noise caused by the
separation. However, the harmful influence of the noise can be eliminated
effectively by providing, as shown in FIG. 9 a protective circuit
of diodes D.sub.1 D.sub.2 and a resistor R.sub.22 in the input
of the rear stage 20b. Moreover, as shown in FIG. 10 a field indication
type vortex flow meter can be accomplished by installing a power
source 40 in the vicinity of the rear stage 20b and driving an indicator
60 by the output of the amplifier 26 in the rear stage 20b. Furthermore,
it is also possible to transmit the pulse signal P.sub.1 from the
front stage 20a to the receiver or receiving side by omissing the
rear stage 20b, as shown in FIG. 11. In this case, the converter
20 and the receiver are connected to each other by three transmission
lines 30 and the pulse signal P.sub.1 is integrated by means of
an integrator 70.
Since other modifications and changes varied to fit particular
operating requirements and environments will be apparent to those
skilled in the art, this invention is not considered limited to
the examples chosen for purposes of disclosure, and covers all changes
and modifications which do not constitute departures from the true
spirit and scope of this invention . |